A radio receiver may be used to recover a “baseband” signal (e.g., a radio signal having a first frequency) from transmitted data (e.g., typically having a second frequency different from, and oftentimes higher than, the first frequency). In some cases, the baseband signal may include frequencies near 0 Hz (e.g., 1 Hz). In some wireless communication signal systems, transmitted signals can include original low frequency radio signal portions that are modulated to the higher transmitted carrier frequencies (e.g., in a radio-frequency [RF] signal) for transmission. Such original low frequency components (i.e., the baseband radio signal) can then be converted or recovered from the relatively high frequency components by using a radio receiver.
In a typical conversion to baseband signal frequencies, one or two mixers or multiplier circuits can be used for a “direct down” conversion approach where incoming data (e.g., a radio signal) is directly converted from the transmission frequency or broadcast channel (e.g., typically from about 40 to about 60 kHz) to the baseband frequency (e.g., about 1 Hz) in a receiver. However, one drawback of this approach is a potential mismatch between the transmission frequency and a reference frequency of the receiver. If those frequencies are not identical, a “delta” frequency or frequency difference will be converted into a signal portion or component in a receiver output waveform for the recovered radio signal.
A radio signal may generally be sinusoidal in nature, and may be made up of different components. For example, if a transmitted signal has an AM sinusoidal waveform (e.g., generally having the form A(t)cos(ωct), where A(t) can be any time-varying signal, cosine may indicate a sinusoidal waveform, ω may represent an angular frequency, where ω=2πf, and “f” is the frequency of the sinusoidal waveform, and “t” can represent time), and the receiver has a reference oscillator providing a sinusoidal, or ideally, a square wave signal (e.g., generally having the form B cos(ω1t+φ) plus its odd harmonics, but simplified to the fundamental frequency discussion herein), a mixer output can be defined as A(t)B cos(ωct)cos(ω1t+φ), which equals ½A(t)B[ cos((ω1+ωc)t+φ)+cos((ω1−ωc)t+φ)]. Also, when the sum frequency term (e.g., B cos((ω1+ωc)t+φ)) is removed by a low-pass filter, this mixer output can be simplified further, as shown below in Equation 1:Mixer Output=½A(t)B cos((ω1−ωc)t+φ)=½A(t)B cos(Δωt+φ)  (1)
As shown, A(t) is the desired recovered signal and the cos(Δωt+φ) portion is a time-varying signal that may be mistaken for a desired signal, depending on the frequency and strength of this signal portion relative to the signal A(t). Another undesirable result, as reflected in Equation 1, involves the phase error, where the mixer output may be zero when the frequencies of the reference and received signals are identical, and the phase difference is 90°, for example. One conventional solution to this problem is to use a Costas receiver, as shown in FIG. 1, and indicated by the general reference character 100. In this approach, an output of voltage-controlled oscillator (VCO) 108 can be applied to mixer or product modulator 102-I, and also can pass via 90° phase shifter 106 to product modulator 102-Q. In the I-channel, an output of product modulator 102-I can connect to low-pass filter 104-I, the output of which can be applied to phase discriminator 110. Similarly, in the Q-channel, an output of product modulator 102-Q can connect to low-pass filter 104-Q, and the output of which can be applied to phase discriminator 110. Further, an output of phase discriminator 110 can be applied to VCO 108 as a control voltage to form a phase-locked loop (PLL).
When feedback in the PLL is correct, it can force cos(φ)=1, or φ=0 degrees. In this type of architecture, the VCO output frequency may need to be relatively close to the carrier frequency for phase lock. Typically, a crystal can be used to provide such an appropriate VCO output frequency. However, crystal oscillators may not be practical at some transmission frequencies. Thus, a second PLL can be employed with a replica VCO, as shown in FIG. 2, and indicated by the general reference character 200. In FIG. 2, reference 202 can provide a reference clock to PLL 204, which can also receive an output from VCO 208-1. An output of PLL 204 can be input to VCO 208-1 and to summation circuit 210, which can also receive an input from phase discriminator 206. An output from summation circuit 210 can be provided to VCO 208-0, an output of which can be sent to mixers (e.g., product modulators 102-I and 102-Q of FIG. 1). Here, a sum of two VCO control voltages can be used to adjust an oscillator (e.g., VCO 208-0) frequency, where the control voltage from PLL 204 can provide a coarse frequency adjustment, and phase discriminator 206 can provide a fine frequency adjustment along with a phase correction.
One drawback of conventional approaches, such as those shown in FIGS. 1 and 2, is the limited synchronization range of the reference clock to the incoming radio signal. Therefore, a reliable approach for synchronization of a reference clock to a radio signal during baseband frequency recovery over a wider range of frequency differentials is desired.